Transformer-Less Converter Concept

Transformer-less converter conceptfor a grid-connection of thin-film photovoltaic modules Ulrich Boeke Philips Research Laboratories Philips Technologie GmbH Aachen, Germany [email protected] Heinz van der Broeck Institute of Automation Engineering University of Applied Science Cologne, Germany Abstract— A transformer-less converter concept for grid- connected photovoltaic systems is proposed that combines a DC/DC converter front-end with a DC/AC inverter. The converter system has an earth-connected DC input, as required from many thin-film photovoltaic modules. The DC/DC converter increases the positive photovoltaic DC-bus voltage by its negative DC output voltage to supply a grid-connected 3-phase inverter. This architecture extends today’s power electronic converter topologies for thin-film photovoltaic modules considering their special requirements with the ambition to realize higher power conversion efficiencies at lower cost. Thin-film photovoltaic, DC-DC, DC-AC, converter, earth connection, efficiency I. INTRODUCTION Photovoltaic power systems are discussed to contribute in a mix with other renewable energy sources to the energy supply of the 21 st century. However, costs of electricity from photovoltaic modules are still higher compared with utility bulk power especially due to high prices of photovoltaic modules. A significant cost reduction of photovoltaic modules is expected especially from a mass production of thin-film photovoltaic modules that require less, however still complex, production steps than modules with crystalline solar cells [1]. The most interesting application of photovoltaic (PV) power systems are grid-connected power systems. In a mix with other grid-connected renewable energy technologies such as wind or biomass power plants, photovoltaic power systems generate electricity especially in summer times when wind is typically low and grids are often loaded additionally by air- conditioning and other cooling systems [2]. Grid connected photovoltaic systems require DC/AC converters to feed electricity form a DC bus of photovoltaic modules into an AC utility grid. A converter has to load photovoltaic modules with an optimum impedance to operate modules in their maximum power point (MPP). It is important that a converter does not generate AC voltages between the photovoltaic module DC bus and earth to avoid earth-leakage currents. Photovoltaic modules and also a converter in a grounded cabinet may have significant parasitic capacitances to earth. Today’s most efficient photovoltaic converters for crystalline solar cell modules operate with high European Efficiencies of up to 97 % [3]. The term “European Efficiency” is explained in the Appendix. Converters for thin-film photovoltaic systems have, on the other side, an additional requirement that limit European Efficiencies of state of the art converters to 95%. This requirement is an earth-connection of a photovoltaic DC-bus minus-pole to avoid negative electric fields in the photovoltaic modules to earth. That is discussed to avoid a transport of positive charged sodium ions from a module front glass into the transparent conductive oxide (TCO) layer of a thin-film module [4, 5]. Due to these requirements, such a DC-bus system is not able to supply directly a transformer-less, voltage-feed, grid-connected 3-phase inverter. Practical, DC/AC converter systems for thin-film photovoltaic systems are realized by combing either a DC/DC converter front-end with high frequency transformer with a grid-connected DC/AC inverter [6, 7] or by combining a DC/AC inverter with a grid frequency transformer depicted in Figure 1 [8]. Thus, these converters include two converter stages and both low- frequency or high-frequency transformer stages reduce the converter efficiency by at least 2 % compared with transformer-less single-stage inverters for crystalline solar cells. Single-stage DC/AC converters for thin-film module applications without a transformer are also known that make use of the “flying inductor” concept [9, 10]. Products with this technology realize European efficiencies of 95 % too [11, 12]. To extend power electronic converter topologies for thin-film photovoltaic systems, this paper presents a transformer-less converter concept that promises a realization of higher conversion efficiencies. The basic idea of the proposed converter concept is to use a transformer-less DC/DC converter to increase a positive photovoltaic DC-bus voltage of e.g. +500 V by its output voltage of e.g. –350 V to a suited DC input voltage of a grid-connected 3-phase inverter illustrated with Figure 2. ~ V DC + - N L V AC Grid frequency transformer Q 1 Q 2 Q 3 Q 4 L 1 L 2 M C 1 C 2 L 3 PV Figure 1: Typical circuit diagram of a photovoltaic inverter with grid frequency transformer 978-1-4244-2279-1/08/$25.00 © 2008 IEEE 1 II. THIN-FILM PHOTOVOLTAIC MODULES A wide range of thin-film photovoltaic module technologies exists today that make use of different chemical composites. The proposed converter concept requires thin-film modules that are approved to realize DC bus voltages of at least 700 V. Table III in the Appendix documents modules that fulfill this requirement. A proper interaction of modules and converter requires the consideration of a module voltage range that is a function of temperature, solar radiation and load. A mathematical model to calculate that voltage range has been proposed in [13]. It considers the first module in Table III from Wuerth Solar as an example. For other modules in Table II it is estimated that they operate with output voltage between 55 % and 100 % of V oc (-10°C) too. The index “oc” stands for open-cathode as an acronym for an unloaded solar cell. V oc (-10°C) is often considered as worst case operation point where a photovoltaic module generates its maximum output voltage. The last column in Table III informs about the number of series connected modules per string that generate the required DC input voltage range for the proposed converter. These numbers of modules in series result first in MPP-voltages between 350 V…550 V and second DC-bus voltages of unloaded modules of maximum 700 V. Multiple strings can be connected in parallel to design systems with different nominal power levels. New thin-film modules using amorphous silicon technology degrade in the first weeks of operation. Manufacturers offer this as free over-power. That also increases the maximum module voltage, V oc up to 11 %. Thus it is preferred to install such photovoltaic systems in summer periods when module operating temperatures are not too low. III. CONVERTER CONCEPT The proposed converter concept consists of a DC/DC converter and a 3-phase DC/AC inverter depicted in Figure 2. The earth-connected DC voltage of a photovoltaic module string is the positive DC supply voltage of the inverter. Its value can change between a value slightly higher than the peak voltage of an AC line-to-ground voltage and a certain maximum value. The DC/DC buck-boost converter generates from the positive DC input voltage V in = V DC.1 a second constant DC voltage V DC.2 that is the negative supply voltage for the inverter. That is similar to the converter concept in [14]. That converter concept, however, considers that PV- modules also generate the earth-connected negative DC supply voltage of an inverter that is not recommended for the use of thin-film PV modules as discussed above. Thus the converter concept in this paper extends the converter concept from [14] to fulfill the special requirements of thin-film PV-modules. Vin +350 V.... +700 V . PV C4 C5 L1 L3 VDC.2 +350 V Control DC/DC Converter DC/AC Inverter L4 C7 C8 C9 400 V 3-phase AC grid Vinverter Q1 D1 D3 Q3 Q4 Q5 Q7 Q6 Q8 D9 D4 D6 D5 D7 D8 Iplus Iminus Figure 2: Principle circuit diagram of the proposed converter system A 3-phase inverter has been chosen for the converter concept because of the lower current stress in buffer capacitors C4 and C5. Transformer-less 3-phase inverters for photovoltaic applications with crystalline solar cells and high efficiencies up to 98 % are known [15, 16]. Thus, the paper focuses in the following on the design of an efficient DC/DC buck-boost converter for the proposed converter concept. After that the operation of voltage-feed 3-phase inverter with two unbalanced DC supply voltages will be discussed. A. DC/DC Converter The converter system benefits from a realization of a DC- DC converter that operates between zero and full-load with very high efficiency. Thus the use of a soft-switching active- clamped buck-boost converter is proposed that is depicted in Figure 3. The active clamping mechanism is realized by means of additional components switch Q2 with inverse diode D2 and snubber capacitor C2, clamping capacitor C6 as well as inductor L2. Classical PWM-controlled active-clamped DC/DC converters operate with fixed switching frequency [17, 18]. Hereby, the clamping voltage changes as a function of the relative power level and relative input to output voltage transfer ratio. One challenge of an active-clamped buck-boost converter design in the given application is a limitation of the clamping voltage V clamp to a relative small value. A large clamping voltage results either in a low useful converter DC input voltage range or an unfavorable high voltage stress of the power semiconductors. To limit the clamping voltage a different control principle has been chosen for the active- clamped buck-boost converter in this paper. The buck-boost converter operates with two independent and constant regulated DC voltages V DC.2 and V clamp at medium to maximum power levels. A variable switching frequency is a consequence of that operation “Mode 2”. Below a certain medium power level the converter is forced to operate with its maximum specified switching frequency in “Mode 1”. The clamping voltage is no longer regulated and drops below its regulated value of “Mode 2” that has been learned from an analytical converter model. In “Mode 1” the converter behavior has a floating transition from an active- clamping converter to a bi-directional soft-switching buck- boost converter similar to the resonant-pole principle [19]. C4 C5 L1 L2 C6 C3 C1 C2 D3 D2 D1 Q1 Q2 IL1 IL2 Vclamp Vinveter PV Vin VDC.2 Figure 3: Principle circuit diagram of an active-clamped buck-boost converter 2 To study the converter behavior two converter models have been developed. An analytical converter model has been used to study the influence of the control principle with its switching frequency limitation in Mode 1 and regulated clamping voltage in Mode 2. That analytical model considers a loss-less, hard-switching converter. The second more complex converter model includes soft- switching details e.g. capacitors C1, C2, C3, dead-time intervals between gate signals of Q1 and Q2 and current time functions in L1 and L2 during these dead time intervals. This second model can be calculated, however, only numerically. All details of the more complex model are considered in the converter time functions depicted in Figure 4 that considers data of the prototype documented in Chapter IV. A practical difference between both models is that the complex model considers a current drop in inductor L2 during the dead time after the turn-off of power semiconductor Q1. Due to that I L2 (t a +t b ) is slightly smaller than –I L2 (t .a ) in opposite to the definition of equation (9). Time interval t b is slightly shorter and the switching frequency is slightly higher than calculate with the analytical model. 0 10 20 30 40 50 60 time (µs) Figure 4: Numerical calculated time functions of an active-clamped buck-boost converter Vin = +350 V, VDC.2 = +350 V, Vclamp = +50 V, P = 3 kW, fs = 19.3 kHz 1) Analytical DC/DC converter model The analytical converter model of a loss-less, hard- switching, active-clamped buck-boost converter considers a set of nine equations that includes: • Difference equations (1), (2), (3), (4) of the effective inductance in four time intervals t a , t b , t c , t b +t c of a switching frequency period, • The definition of the switching frequency (5), • The definition of input and output power (6), (7), (8), • A condition of the active-clamping mechanism to operate with constant average charge in a clamping capacitor (9). a t in V 2 L 1 L 1 I 2 ⋅ ⋅ + = ∆ ⋅ (1) [ ] b clamp 0 t V 2 L 1 I I 2 ⋅ ⋅ = ∆ + ⋅ (2) [ ] c 2 . DC in 0 t V V 2 L 1 I 2 ⋅ + ⋅ = ⋅ (3) ( ) c b 2 . DC t t V 1 L 1 I 2 + ⋅ ⋅ = ∆ ⋅ (4) c t b t a t s f 1 + + = (5) [ ] s c s a 0 in in f t I f t I V P ⋅ ⋅ ∆ − ⋅ ⋅ ⋅ = (6) ( ) [ ] s c s c b 0 2 . DC out f t I f t t I V P ⋅ ⋅ ∆ + ⋅ + ⋅ ⋅ = (7) P in = P out = P (8) ( ) ( ) a t 2 L I b t a t 2 L I − = + (9) In Mode 1, these equations have been solved to calculated the parameters ∆I, I, t a , t b , t c , V clamp . 2 . DC 2 1 in 1 2 . DC in max . s V ) L L ( V L V V f 2 1 I ⋅ + + ⋅ ⋅ ⋅ ⋅ = ∆ (10) 2 . DC in 2 in 2 1 in max . s 0 V V L V L L 1 V f I 2 P I + − + ⋅ ⋅ ⋅ ∆ ⋅ = (11) in 2 1 a V I 2 ) L L ( t ∆ ⋅ ⋅ + = (12) 2 . DC in 0 2 c V V I 2 L t + ⋅ ⋅ = (13) c a max . s b t t f 1 t − − = (14) b 0 2 clamp t ) I I ( 2 L V ∆ + ⋅ ⋅ = (15) In Mode 2, these equations are used to calculate parameters f s , I, ∆I, t a , t b , t c as well as a current ratio A. Figure 5 till Figure 7 illustrate data calculated with parameters of the prototype documented in Chapter IV. On Off VGate(Q1) VGate(Q1) VGate(Q2) VCE(Q1) V(D3) V(D3) I(L1) I(L2) -I(D3) 800 V 600 V 400 V 200 V 0 V 30 A 20 A 10 A 0 A -10 A -20 A -30 A -40 A t a t b t c I 0 ∆I ∆I 3 clamp 2 . DC 2 1 2 . DC in 2 . DC clamp 2 . DC 2 V V L L V V V V V L A − | | . | \ | + + = (16) ( ( ( ( ( ¸ ( ¸ + ⋅ ⋅ ⋅ ⋅ ⋅ + + ⋅ ⋅ ⋅ + + + ⋅ ⋅ ⋅ + ⋅ ⋅ + ⋅ ⋅ + − + = ) V V ( V V A V V L ) V V ( V L ) A 1 ( ) V V ( V V A ) V V ( V ) L L ( A ....... V V L V L L P I 2 . DC in 2 in clamp clamp in 2 2 . DC in in 2 2 . DC in 2 in clamp 2 . DC in clamp 2 1 2 . DC in 2 in 2 1 0 (17) 0 I A I ⋅ = ∆ (18) in 2 1 a V I 2 ) L L ( t ∆ ⋅ ⋅ + = (12) clamp 0 2 b V ) I I ( 2 L t ∆ + ⋅ ⋅ = (19) 2 . DC in 0 2 c V V I 2 L t + ⋅ ⋅ = (13) Figure 5 till Figure 7 illustrate calculated operation parameters considering data of the prototype that is discussed in Chapter IV. The transition between Mode 1 and Mode 2 is at about 1200 W when operating with V in = 350 V and at about 1750 W when operating with V in = 550 V. The switching frequency is limited to 50 kHz in Mode 1. In Mode 2 the clamping voltage is regulated to 50 V and the switching frequency varies between 19 kHz and 50 kHz as function of input voltage and power level. Figure 7 illustrates that the converter operates also at no- load with a peak current of 3.4 A in L2 at the time points when Q1 and Q2 are turned off. That allows soft- switching even at no-load. Switching frequency (kHz) 0 1000 2000 3000 0 10 20 30 40 50 60 Power (W) Figure 5: Calculated switching frequencies Clamping voltage (V) 0 1000 2000 3000 0 10 20 30 40 50 60 Power (W) Figure 6: Calculated clamping voltage Inductor peak current (A) 0 1000 2000 3000 0 5 10 15 20 25 Power (W) Figure 7: Calculated inductor peak current in L1 and L2 Vin = 550 V Vin = 350 V Vin = 550 V Vin = 350 V Vin = 350 V Vin = 550 V 4 2) DC/DC converter control The controller sub-circuit of the active-clamped buck- boost converter, depicted in Figure 8, makes use of two PI- type voltage regulators and a dual-peak current control circuit. The voltage controller of the buck-boost converter output voltage, V DC.2 generates the reference signal for the positive peak currents in inductors L1 and L2. Once the regulated current and energy levels in L1 and L2 has been reached comparator 2 generates a reset signal for a flip-flop that turns- off Q1 and turns-on active-clamping switch Q2. The voltage controller of the clamping voltage, V clamp generates the reference signal for the negative peak current of the inductor L2. Once this current level has been reached comparator 1 generates the set-signal for the flip-flop that turns-off Q2 and turns-on Q1. Additionally comparator 1 feeds a mono-stable circuit (a one-shot timer) that generates a pulse with an equivalent length equal to the time period of the maximum switching frequency. Thus the next set-signal for the flip-flop can be generated first if the inverted output of the mono-stable circuit (one-shot) is high again. Our converter prototype uses a LEM “LTSR 25-NP” current shunt to monitor the current time function in inductor L2. That current shunt generates with two primary turns an output signal of 0.05 V/A and it has a peak current range of ±40 A that is transferred into output signals of ±2.0 V. The output signals of the two voltage regulators are limited such that the peak current reference signals do not exceed signals equivalent to ±35 A. That is important e.g. during the start of the converter when capacitors C5 and C6 are charged with maximum allowed current to generate the nominal values of V DC.2 and V clamp . The output signals of the flip-flop are feed first into a sub- circuit that generates dead-time signals and second into a standard level-shifter (IR2213) to drive the two power semiconductors Q1 and Q2 of the buck-boost converter. The actual prototype uses constant dead-time signals to support the soft-switching of the converter. That can be principally extended to adapted dead-time generation that monitors the dV/dt of power semiconductor Q2 as given in [18]. IL2 V(IL2) VDC.2 Vclamp One Shot 20us & S R Q Q Gate Q1 Gate Q2 Flip Flop Comp 1 Comp 2 VLimitter VLimitter VRef 2.5 V VRef 2.5 V PI-type voltage regulator 1 PI-type voltage regulator 2 Figure 8: Principle control circuit B. DC/AC Inverter The 3-phase DC to AC inverter shown in Figure 2 consists of 3 independent single phase inverters which are supplied by unsymmetrical DC voltages (V in and V DC.2 ). As illustrated in Figure 9 each inverter feeds an almost sinusoidal current into the grid. This current is set by a closed loop control circuit and pulse width modulation (PWM). The carrier or switching frequency of the PWM is much higher than the grid frequency. g s f 1 f >> τ = (20) Thus, only small filter components are required to sufficiently attenuate all high frequency harmonics of the PWM. As the grid frequency depending voltage drop at the filter components (L f , R L ) is very low, the duty cycle of the PWM: ) τ + τ − ⋅ + ⋅ τ = 2 / t 2 / t 2 . DC in 1 1 dx V V ) x ( V 1 ) t ( V (21) almost follows the AC grid voltage: ( ) t f 2 sin V ˆ ) t ( V g g g ⋅ ⋅ π ⋅ ⋅ = (22) superposed by the lower DC link voltage U DC.2 . Considering the modulation factor 1 V V ˆ m 0 2 . DC g ≤ = < (23) this means the PWM can be described by the equation ) t f 2 sin( m V V V ) t ( V g 2 . DC in 2 . DC 1 ⋅ ⋅ π ⋅ ⋅ + + ≈ . (24) As the grid current I g (t) is controlled offset free, the average currents supplied from the two different DC voltages for all three inverter legs are equal. 2 . DC in inverter 2 1 V V P I I + = = (25) Moreover, the power flow in symmetrical 3-phase system is constant so that the DC supply voltages are not stressed by low frequency harmonics. Vin Vdc2 Q4 Q3 D3 D4 ~ Vg Ig Lf RL V1 I1 I1 Figure 9: Principle circuit diagram of one inverter phase D9 I 2 I 1 V in V DC.2 V g V 1 I g 5 The power to be supplied by the buck-boost converter is equal or less than half the inverter output power. 2 . DC in 2 . DC inverter boost buck V V V P P + ⋅ = − (26) The operation of the 3-phase inverter has been studied by means of a MathCAD simulation. For illustration purposes the frequency ratio considered in the simulation is low: f s /f g = 33. In Figure 10 the PWM generation of all three phases is shown based on the highest voltage ratio V in = 2*V DC.2 . In addition, the normalised PWM voltage V 1 (t) of phase “R” is shown in Figure 11. The corresponding grid current I g (t) and the overall switched inverter input current I 1 (t) are depicted in Figure 12. PWM input signals 0 0.33 0.67 1 0 4 8 12 16 20 t (ms) Figure 10: PWM carrier and inverter output reference signals Vin = +700 V, VDC.2 = +350 V, fg = 50Hz, fs = 1.65 kHz Normalized PWM voltage V1(t)/[Vin+VDC.2] N 0 0.5 1 0 4 8 12 16 20 t (ms) Figure 11: Calculated normalized PWM signal of phase “R” Vin = +700 V, VDC.2 =+ 350 V, fg = 50Hz, fs = 1.65 kHz Normalized inverter currents N 1 0 1 0 4 8 12 16 20 t (ms) Figure 12: Calculated inverter input current I1 and grid current Ig of phase “R” IV. EXPERIMENTS An active-clamped buck-boost converter prototype has been build to verify the calculated converter functionality. Parameters of the converter depicted in Figure 3 are collected in Table I. The key components of the active-clamped buck-boost converter design are the two inductors L1 and L2. The value of active-clamping inductor L2 is a compromise of stored energy to support soft-switching and the switching frequency range. An inductor with less than 25 µH does not store enough energy to support the soft-switching of Q1 and Q2. An inductance of more than 25 µH results in a lower minimum switching frequency than 19 kHz that could be optionally compensated with a higher clamping voltage. The inductance value of L1 has been identified with the diagram in Figure 13. The stored energy in L1 has a minimum for L1= 500…600µH. An inductance value of L1 = 600 µH has been chosen since a lower value results in larger current stress and lower minimum switching frequency than 19 kHz too. Inductor L1 has been realised by means of a four inductor components in parallel. TABLE I. BUCK-BOOST CONVERTER PROTOTYPE DATA Parameter or Component Value Vin 350 V…700 V VDC.2 350 V Vclamp 50 V P 0 W…3000 W fs.max 50 kHz C1, C2 output capacitances of Q1+D1, Q2+D2 C3 output capacitance of D3 C4, C5, C6 680 µF L1 600 µH, 4x E65-3C94 L2 25 µH, ETD49-3C94 Q1, Q2 Infineon IHW25N120R2 D3 Cree C2D10120A Energy (mJ) 0 50 100 150 200 250 0 200 400 600 800 1000 L 1 (µH) Figure 13: Calculated peak energy values in inductors (L2 = 25 µH) E L1.peak 10*E L2.peak Phase R Phase S Phase T I g I 1 0 6 Measured time functions of an active-clamped buck-boost converter prototype are depicted in Figure 14. The converter operates in this operation point in Mode 1 with 20% of its nominal power level and an efficiency of 90 %. The collector- emitter voltage of the IGBTs is nicely clamped to 731 V due to the active-clamping mechanism. The control circuit generates two gate signals that have fixed dead time intervals of about 400 ns. That results in a dV/dt of about 731V/500ns at the IGBTs. Hereby, only the output capacitances of semiconductors Q1+D1, Q2+D2 and D3 are used as snubber capacitances C1, C2 and C3. The measurement also shows an unwanted oscillation of the inductor current I L2 after the turn-off of IGTB Q 2 . Thus it is difficult to identify the end of time period t c from that measurement that is an issue of further investigations. Figure 14: Measured buck-boost converter time functions Vgate = 10 V/div, VCE = 200 V/div, IL2 = 5 A/div, t = 5 µs/div Vin = +350 V, Vout = +358 V, Vclamp = +23 V, Pin = 660 W, Pout = 596 W V. SUMMARY A transformer-less converter concept for a grid-connection of thin-film photovoltaic modules is proposed. The converter concept uses a DC/DC converter to increase a positive and earth-connected photovoltaic DC-bus voltage by its negative DC output voltage to supply a grid-connected DC/AC inverter. Since the DC/DC converter processes only 50 % of the power of the inverter, the concept describes a quasi 1.5-stage converter. An active-clamped buck-boost converter is proposed as DC/DC converter since it offers soft-switching operation with moderate internal reactive power flow. The control of the buck-boost converter considers an operation in two modes either with limited maximum switching frequency in Mode 1 at low power levels or with regulated clamping voltage in Mode 2 at medium to maximum power levels. VI. OUTLOOK The authors would like to motivate a debate weather DC/AC inverters for grid-connected photovoltaic systems must have an insulation transformer that are currently required by legal standards in some countries or by manufacturers of thin-film photovoltaic modules. As alternative, only an earth- connect DC-bus voltage is proposed as an alternative specification. That can be realized with less effort compared with high- or grid-frequency transformers in photovoltaic converters as proposed in this paper as well as in [9, 10]. ACKNOWLEDGMENT The authors thank Mr. Jost Wendt from Infineon for the support with IGBTs and the discussion of their use in soft- switching power converters. IL2 VCE VQ1.gate VQ2.gate ta tb tc 0V 0V 0A 7 APENDIX A. European Efficiency The “European Efficiency” value η Eu of converters for photovoltaic applications, proposed by R. Hotopp, is a definition of an average efficiency value that considers different weighting factors for six different power levels to consider the probability of different solar radiation levels in Europe [1]. TABLE II. PARAMETERS TO CALCULATE THE “EUROPEAN EFFICIENCY” Relative power level 5 % 10 % 20 % 30 % 50 % 100 % Weighting factors 3 % 6 % 13 % 10 % 48 % 20 % %) 100 ( P %) 50 ( P %) 30 ( P %) 20 ( P %) 10 ( P %) 5 ( P Eu 2 . 0 48 . 0 1 . 0 13 . 0 06 . 0 03 . 0 η ⋅ + η ⋅ + η ⋅ + η ⋅ + η ⋅ + η ⋅ = η (20) B. Thin-film photovoltaic module data Data of available thin-film photovoltaic modules are collected in Table III. These modules fulfil the specification of a DC-bus system voltage of at least 700 V. A PV-module voltage in the maximum power point (MPP) V MPP (+25°C) is given in data sheets. A PV-module voltage of an unloaded module (oc = open-cathode) at -10°C is used typically as maximum considered module voltage. It can be calculated by using V oc (+25°C) and its temperature coefficient that are both given in module data sheets. The minimum PV-module MPP- voltage occurs at its maximum operating temperature. Typical +70°C are considered at maximum operating temperature. The minimum MPP voltages in Table III are estimated to be 55 % of V oc (-10°C) based on the solar cell model in [13]. Further abbreviations in Table II are explained below. a-Si Amorphous silicon thin-film technology CdTe Cadmium-Telluride thin-film technology CIS Cooper-Indium-Disolenoid thin-film technology CIGS Cooper-Indium-Germanium-Disolenoid thin-film technology Wp Watt-peak – solar module power level at standard test conditions µ-Si Microcrystalline silicon thin-film technology TABLE III. DATA OF THIN-FILM PHOTOVOLTAIC MODULES SORTED BY EFFICIENCY TAKEN FROM DATA SHEETS IMPP, VMPP, Voc at a solar radiation of 1000 W/m 2 and air mass = 1.5 Manufacturer Tech- nology Thin-film module type Nominal power Module Efficiency IMPP (25°C) VMPP(+70°C) ~Voc(-10°C)/1.8 VMPP (+25°C) Voc (-10°C) Modules per string Units Wp % A V V V Wuerth Solar CIGS WSG 11007/80 80 11.0 2.23 ~31.2 35.9 50.1 13…14 First Solar CdTe FS-272 72.5 10.1 1.07 ~54.4 67.9 97.9 7 Sulfurcell CIS SCG60-HV 60 7.3 1.48 ~31.8 41 57.3 12 CSG Solar GmbH µ-Si CSG-90 90 6.5 1.45 ~53.1 63 95.7 7 United Solar Ovonic a-Si ES-127 127 6.5 4.2 ~29.5 30.2 53.1 12…13 Schott ASI 86 a-Si ASI 86 86 5.9 5.49 ~14.4 19.1 26 25…26 REFERENCES [1] H. Haeberlin: Photovoltaik, VDE Verlag, ISBN 978-3-8007-3003-2, 2007, in German [2] The combined power plant project: www.kombikraftwerk.de [3] M. Kaempfer, U. Zwahlen, H. 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