IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 38, NO.2, APRIL 1991 135 Design of a High-Frequency Planar Power Transformer in Multilayer Technology Dirk van der Linde, Corlex A. M. Boon, and J. Ben Klaassens Abstract-A high-frequency power transformer in multilayer printed circuit board (ML-PCB) technology is presented for applications in switched-mode power supplies operating at frequencies up to several megahertz. The mechanical configuration of laboratory prototypes is discussed, as well as the electrical, parasitic, and thermal behavior. The presentation is focused on the leakage inductance since the analysis of other aspects is relatively simple. Test results show high efficiency, low leakage inductance, good thermal behavior, and line insulation properties of the transformer. Further, the topology enables the designer to make a tradeoff between leakage inductance and interwinding capacitance. Due to the well-defined geometry, parasitic interwinding capacitance and leakage inductance are reproducable and can be computed relatively easily. I. INTRODUCTION I N THE NEW generation of power converters, miniaturization has become an important rule of design [ 11, [2], [5], [6], [ 111. Modem semiconductors allow fast switching and can be used to increase switching frequencies up to the megahertz region. Consequently, capacitive and magnetic components can be reduced in weight and size. This miniaturization, however, gives rise to some specific problems: 0 The parasitic components set limitations to the hf cut-off frequency. The parasitic behavior of conventional transformers is not reproducable. Sufficient cooling of compact power devices is often a problem. Parasitic components play an important role in circuit behavior due to the high switching frequencies. For good hf properties, the leakage inductance and the interwinding capacitance have to be small; they both limit the hf cut-off frequency [4]. Energy stored in the parasitic leakage inductance may result in high-voltage peaks during switching of the vulnerable switching devices. These peaks cause dynamic power loss and excessive stress on components. A small leakage inductance can be achieved by a good inductive coupling between the primary and secondary. In practice, the windings are interleaved with small distances between the windings. This leads, however, to a high interwinding capacity. The requirements for a low leakage inductance and a low interwinding capacitance are contradictory. Therefore, the “LC product” of a transformer can be used as a “figure of trouble.” These parasitic effects and the related hf properties in conventionally wound transformers appear to be uncontrollable [7]. The Manuscript received February 20, 1990; revised December 1990. D. van der Linde and C. A. M. Boon are with Hollandse Signaalapparaten B.V., the Hague, the Netherlands. J . B. Klaassens is with Delft University of Technology, Delft, the Netherlands. IEEE Log Number 9142807. variables in the manufacturing process cause considerable tolerances in the winding geometry. A high degree of reproducibility is fully related to a strictly defined winding geometry. Only if reproduction can be guaranteed, an attempt is useful to calculate parasitic components. A compact converter module has a relatively small surface area for the conveyance of internally dissipated heat. To keep hot-spot temperature rise under control, a high converter efficiency is required. Additionally, a flat package provides the largest surface area and, therefore, the best heat transfer to the environment. For this reason, there is a tendency towards converter modules in flat-pack housing. The shape of a conventional transformer, however, is not very suitable for use in flat-pack modules. In order to approach the problems mentioned above, the idea was to design a transformer with its windings integrated into a multilayer printed circuit board (ML-PCB). Ferrite core-halves on either side of the multilayer winding package would complete the magnetic circuit (without air gap). This configuration would provide the following advantages. The winding geometry and its related parasitic behavior are defined within the (small) tolerances of PCB manufacturing and are therefore reproducable. Further, the entire transformer can be flat (planar transformer) since the windings consist of thin copper layers. These thin copper layers reduce skin effect losses. The flat configuration provides a relatively large surface area for the transfer of dissipated heat to the environment. The manufacturing process can be fully automated, although multilayer manufacturing requires dedicated facilities and skills. The ML-PCB transformer can be an integrated part of a circuit board with other components. In the following, the configuration of the ML-PCB transformer will be discussed. Then, aspects such as parasitics, line insulation, and thermal behavior will be discussed on the basis of prototypes. 11. MULTILAYER TRANSFORMER A . Windings A multilayer is composed of several double-sided printed circuit boards (bilayers) pressed and cemented together with epoxy resin. Each bilayer consists of a standard epoxy base-layer of 76 pm covered on both sides with a 60-pm copper layer. The epoxy resin layer between the bilayers measures 200 pm. In an ML-PCB, several distinct copper layers are available for creating the transformer windings. To connect individual turns in series or in parallel, interconnections between different layers have to be made by “via holes.” The copper layers intersected by the same via hole are electrically tied together. Fig. 1 shows the interconnection of individual turns in series using via holes. As shown, each copper layer contains one single turn with 01991 IEEE 0278-0046/91/0400-0135$01.00 one should realize that in conventional transformers. = 70 mm V. the primary and secondary windings are split up in two separate groups. Winding arrangement primary. T. In type 1. whereas the window of available standard cores is relatively high. A reduction in the number of flaps needed is achieved using a slightly different approach. using . thus. 38. = 13 900 mm3 m e = 74 gr /L . In this configuration. NO. for hf power applications. the ML-PCB prototypes have an effective window utilization of 0. 3. three types of the ML-PCB transformer were made. as is indicated in Fig. These versions are explained with reference to Fig. two “connection flaps. all via holes are made in one run after multilayer assembly. (B. Although more expensive. = 198 mm2 I . = 330 mT. In type 3-. I Primary Secondary o type 1 Fig. The core should provide openings large enough to enter and leave the winding space with the connection flaps. 2. > 200 “ C . The primary winding consists of eight turns in series.” Flaps to be interconnected are placed such that they can be intersected by the same via hole. 2.PloSs 150 W/dm3 5 at 400 kHz. Alternative winding arrangement primary. secondary windings. each has a different leakage inductance and intenvinding capacitance. A standard core can be adapted to the ML-PCB winding package by grinding off part of each core half. Ferrite Core The selection of a suitable core is another point of discussion. 1 . resulting in a comparable window utilization. The RM-14 is also available in Philips’ new 3F3 femte especially designed for hf power applications. 4. type 2 type 3 Arrangement of windings in prototype. litz wire is mandatory to eliminate the skin effect. On the other hand. 4. 3.3. the primary and secondary windings are interleaved. This application required a 1:8 transformer turns ratio and a center tap on the secondary side (see Fig. This can be realized very easily with the standard via holes. The unmodified RM-14 core parameters are as follows: Effective cross-sectional area effective magnetic path length effective volume of the core mass of the core set relative permeability A . hence reducing the height of the transformer core. 2. turns are connected in parallel. In type 2. APRIL 1991 Fig. Each version differs from the others in winding configuration. whereas in the previous configuration.136 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS. The amount of turns that can be placed in series is limited by the available space for the connection flaps and is determined by the core type. Fig. the limited amount of space for the connection flaps may leave no alternative. the primary winding is “sandwiched” between two groups of B. To study the transformer’s parasitic effects.. = 2000. 50 mT). The ML-PCB transformer’s configuration is flat. Fig. the RM-14 core (Philips) is selected. 3). The epoxy layers occupy a considerable part of the transformer’s winding space. the two turns on each side of a bilayer are placed in series by a “local via hole” or “buried hole. The RM-14 encloses a considerable part of the winding space while sufficient space for the interconnection flaps remains. Winding arrangement secondary.” This requires the special treatment of making via holes in each individual bilayer before multilayer assembly. On “the low side” of the transformer. VOL. The prototype of the ML-PCB transformer was tested in existing lab model 5-V-25-A converters operating at 1 MHz. For the prototypes. as is depicted in Fig. In this case. the energy stored in the magnetic field can be obtained simply by volume integration: E. Roth’s results match the results of measurements on prototypes that are considerably better. 5 mm). The modified RM-14 core has a window that is just high enough to fit the winding package (ca. one needs to study the magnetic field in the transformer’s winding space. 16 T A. A coupling factor smaller than one results in a magnetic field within the winding space of the transformer. In general. the leakage inductance can be computed (formulae (1) and (2)).12 (1) where I is the current through the primary winding. where J is the current density at point ( x . 5 Ij[sin(mia>) - sin(miaj)][sin(nkbj - sin(nkbj)] j=1 ( a ) .. The components B . Roth’s method for describing the magnetic field is very suitable in this case. Twelve bilayers appear to be necessary to obtain the 24 distinct copper layers. Fig.l)a/a n. is the total leakage inductance transformed at the primary side. 4 aA B =--. the configuration was still described as rotation symmetric. Kapp’s formula. 111. Several simplifications are the cause of errors up to 100%. Results obtained by Roth’s method are verified by a two-dimensional finite element analysis using suitable software. 5 mm thick. In the case of the ML-PCB transformers. These methods. a far-too-simple interpretation of Ampere’s law leads to the incorrect conclusion that the leakage flux is restricted to the area between the primary and secondary windings and has the same direction and value in every point of that area. the configuration is described as being rotation symmetric. For each secondary tap. the parasitic behavior of the prototypes will be discussed.VAN DER LINDE et al. = lOab = (k . the core’s window height is reduced by grinding off part of each core half.l)a/b . and L . the originally three-dimensional field analysis can be reduced to a two-dimensional model. by B =- aA ay and E. Within the winding space. however. providing sufficient space for the amount of copper required in power applications. Improvement could be 1 1 . Roth describes the magnetic potential A according to the geometrical parameters as given in Fig. but here. In the case of a (quasi-) two-dimensional configuration A where = i=l k=l 1 A.g. 5: m m In [8] and [lo]. Compared with the outcomes of Kapp’s formula. The core’s weight is reduced by 40 gr and the effective window perimeter by 30 mm. Cross section of the winding area by Roth. a magnetic field can be described by means of a magnetic potential vector A .a j ) (b. Since the multilayer transformer’s windings consist of rectangular copper layers. Once the B components are known. Second. y ) ( i . Therefore. To obtain the value of the leakage inductance. obtained with these simplifications.: DESIGN OF A PLANAR POWER TRANSFORMER I37 the buried-hole method as described with reference to Fig. The magnetic energy E. To adapt the core to the ML-PCB transformer package. From the magnetic field components. precise field calculations were made. showed considerable differences compared with measured values. resulting in a 5-mm-thick ML-PCB winding package.. are known to be imprecise. eight turns are connected in parallel in order to divide copper losses equally between the primary and secondary.. whereas the ML-transformer’s winding package is only ca. and with a relatively flat but wide window. First. 5 shows a radial intersection of an arbitrary transformer configuration. Examined packages either required extensive hardware facilities (mainframes) or gave unsatisfactory results. Now that the potential vector has been determined. y ) . Clearly. Roth describes a radial intersection of the winding space enclosed by ferrite. this approach is insufficient for a satisfactory description of the ML-transformer’s behavior. 2. and B y of the magnetic field can be derived from the magnetic potential A : A . we tried to employ calculation methods known from literature.b j )mink(m f + ni) (3) and where q is the number of individual conductors in the winding space. store in this field is directly related to the leakage inductance L . Next. the components of the magnetic field can be obtained as indicated in formula (4). and I j is the current in conductor j . = iL. describing a radial intersection of the winding space. 5. the results. methods describing the magnetic field inside the winding space of transformers have been presented and have resulted in simple formulae for the leakage inductance. = f p p O j vB 2 d V . . containing rectangular conductors as the windings. mi = cos ( m i x )cos ( n . e. J = 0 (outside the copper) or J # 0 (inside the copper). Roth [ 121 presents analytical solutions to the magnetic field components in quasi two-dimensional structures. PARASITICS HF-BEHAVIOR AND b a. Y ax (4) Fig. (2) A diversity of software packages is available solving these equations by finite element methods. In a second attempt to compute the leakage inductance. 20 mm. representing the parasitic effect of a nonperfect inductive coupling between the primary and secondary windings. The remaining errors are attributed to the inaccuracy of describing the configuration as rotation symmetric. The original RM-14 core has a window height of ca. Leakage Inductance The leakage inductance is a lumped element. The result is a flat transformer configuration. Hence.138 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS. NO. an exchange between the leakage inductance and interwinding capacity can be achieved simply by changing the winding configuration. (c) first-order approximation. = 198 mm2 N 8 Le = 633pH (measured: 640 p H ) C. In Fig. Cps can be computed easily since the windings consist of parallel. It is desirable to keep Cps small. 7(c). it is the price one has to pay for the very low leakage inductance. Table I gives the results with respect to the ML-PCB transformer prototypes. The results are displayed as line elements in the winding space in Fig. a configuration of one primary and one secondary turn is discussed. in Table 11. This transformer with the winding configuration according to type 2 has the lowest “figure of trouble” and is the best approximation of the ideal transformer. the approximation’s main errors . Interwinding Capacitance The parasitic capacitance Cps between the primary and secondary windings strongly affects the transformer’s hf properties. one could hardly expect better results. more complicated. This can be attributed to the use of flat windings at close interwinding distances in combination with the epoxy’s E. A line element shows the direction of the magnetic field at that point of the winding space. According to Table 11. 6. 7(a).1 % of the magnetizing inductance. 2. = 7. The winding space is surrounded by high-permeability ferrite. 38. has an effective window perimeter approximately 2 x 15 = 30 mm smaller than the unmodified core. At every single grid point. and Cps. results were verified with “finite element” software. These lines also provide information related to the direction of the magnetic field at each point of the line but do not give direct information about the absolute value of the magnetic field. The modified RM-14 core. On the other hand. permeability of air relative permeability of the ferrite core A . APRIL 1991 occur around the windings’ edges and in the space between the windings and the ferrite. which is. made by using an accurate three-dimensional (finite-element) analysis. the ML-PCB transformer has a relatively high interwinding capacitance. In this case. the ML-PCB transformer has a very low leakage inductance and a relatively high magnetizing inductance. The formulae according to Roth’s theory are implemented in a straightforward Pascal program for a personal computer (with coprocessor). in this case p r = 2000 (3F3) I. The overall hf-properties are determined by the product of L . the magnetic-field components are computed. the requirements for a low leakage inductance and a low interwinding capacity are contradictory. (b) finite element calculation. 6. (C) Fig. As mentioned. the Cps of 589 pF will be unacceptable in most cases. Cross-section winding area. Nevertheless. With the configuration of flat conductors at close distances in combination with the epoxy’s E. the leakage inductance drops under 0. effective core area le effective window perimeter. just like a compass needle. As shown. = 7. Magnetizing Inductance Fig. In the case of the ML-PCB transformer. Fig. To complete the picture. The magnetizing inductance is determined by the core parameters and by the number of primary turns: where po p. As mentioned before. To illustrate the basic magnetic field inside the multilayer transformer. VOL. The capacitance between two windings can be found simply by using the formula for the capacitance between two parallel conductive plates. the line elements are shown according to the approximation leading to Kapp’s formula. this ML-PCB transformer is compared with a conventional 1:8 transformer on an RM-10 core designed for similar applications and with an extrapolation of the transformer designed by Estrov [ 2 ] . flat conductors. The intersection is divided into an n x m grid. Results of computer simulation of the magnetic field lines in the winding area: (a) Expression following Roth. = 40 mm = A. The line elements do not provide direct information about the absolute value of the magnetic field. of course. Clearly. Secondary B. the capacitance between the windings is not particularly small. In type 3. The radial intersection of the winding space is given in Fig. 7. 7(b) shows the magnetic lines of force according to the MAGGIE software package (Philips). obtained by grinding off 15 mm of its window height. Table I1 shows the parasitics and LC product of a prototype of the planar transformer. Nevertheless. . 8. the designer can trade leakage inductance against interwinding capacitance in a predictable way by changing the arrangement of the windings.. NO. Fe P/2 % I (b) Model heat transfer: (a) Smgle-slded cooling (print-board mounted). . Fig. APRIL 1991 .1 % of the magnetizing inductance. the LC product remains low. the leakage inductance is less than 0.- qq----==(p L1 E1 Fig. and properties are discussed on the basis of prototypes.. Although the interwinding capacitance is relatively high. The use of a multilayer winding package provides enough copper for power applications up to 200 W for the modified RM-14 core..140 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS. 9. CONCLUSIONS A transformer in multilayer printed circuit-board technology is presented. It may be worthwhile to investigate possibilities to reduce these tolerances in the production process. 0 *P'J 0 rc.in copper U0 (f=+=-jq Fig. ______. Experimental transformer. 38. . (b) double-sided cooling VII. Only the tolerance in the epoxy resin layers causes tolerances of 10% in the parasitic leakage inductance and interwinding capacity. The configuration has a low leakage inductance and a high magnetizing inductance. 2. Further. The prototypes of this compact transformer showed good line insulation properties and thermal behavior. VOL. setting some limitations to the reproducibility. 9 shows an experimental transformer that has been tested in a boost converter operating at 1 MHz. 8. requirements are within the thickness of the epoxy resin layers.Only the thickness of the epoxy resin layers is subject to some ence of small air bubbles in the epoxy resin. the ML-PCB In the prototypes. it will be difficult to establish a thermal represent the temperatures in each layer. voltage between primary and secondary still exceeded 4 kV. During multilayer assembly. For the thermal resistance of ally. the The ML-PCB configuration seems very suitable for hf power insulated side of the multilayer will reach 15°C above heatsink applications where a low leakage inductance has priority over a temperature. According to manufactur. Since the between windings and core is less than expected. This tolerance affects the reproduction of the transformer’s can be made by assuming that this is the only mechanism for heat transfer. 975 nH 365 nH 130 nH meas. . hence provid. tolerance.00034 “C/W 0.: DESIGN OF A PLANAR POWER TRANSFORMER 139 TABLE I Type 1 2 3 meas. A tolerance of The main mechanism for the disposal of the dissipated heat is 15% can be expected. 1050 nH 450 nH 285 nH L S calc . Both situations can be determined. 119 pF 159 pF 517 pF LSCPS meas.8 W at ISCDS a frequency of 1 MHz. The multi.transformer can handle high secondary currents. VI.68 S’ s2 s2 the different layers yields TABLE I1 transformer RM-10 Rprim. especially when heat transIf an off-line power supply needs to provide line insulation. THERMAL MANAGEMENT over the thickness of the resin layers is limited. fer can be realized in both directions.9 mO 1800 nH 50 pF 300 pH 90 fs2 60-pm copper layer 76-pm epoxy layer 200-pm resin layer Ferrite (PCB to heatsink) 0. which is perpen.ferrite and the multilayer on both sides since the thickness of the vidually surrounded by epoxy (base-layer) and epoxy resin multilayer is subject to tolerances due to variations in the (between the bilayers). The tolerance on the epoxy base layers and the copper dicular to the layers. In the case of heat transfer to both sides of the low interwinding capacity.74 1.layout of all layers has to remain within small tolerances of PCB layer’s edges probably suffer from very slight damage caused by manufacturing. 640 pH Le calc. the ing insulation up to 6 kV. Although these results are very encouraging for a One of the objectives while choosing the ML-PCB configuracompact transformer under these circumstances. DC R S . it the transformer will have to meet certain insulation require. 10%.5 kV.might be difficult to establish a thermal contact between the ments. in combination for “creepage and clearance” are simply not applicable.distributes 5 V-35 A without running hot.4 mfl 0. the LC product remains low due to the low leakage inductance. As shown. With an equal amount of core loss. 2. In the ML-PCB transformer. the breakdown power. The computed node voltages tolerance in the thickness of the entire multilayer winding package. After prototypes were exposed to a transformer’s power dissipation is close to 1% of the total output 90% humidity at 40°C during a five-day period. IV. the maximum temperature rise in the configurations according to Fig. Therefore. the resin can handle 30 V/pm. resistors representing the thermal resistance of the individual The tolerance in the epoxy resin layers also results in a layers. . CPS 10. the distinct bilayers are cemented together under pressure. In the case of the prototypes. the maximum temperature rise occurs in the middle of the multilayer at 5°C above heatsink temperature. all copper layers are indi. REPRODUCTION whereas breakdown between windings and core occurred at ca. In the laborarated by a 200-pm epoxy resin layer. therefore the capacitive and inductive coupling are not precisely conduction to both sides of the multilayer. a high degree of reproduction was expected. control V . For this situation. 4 are calculated. conduction parasitic behavior since the distances between the windings and of all dissipated heat to one side of the multilayer and second. as is indicated in Fig.only.7 mQ per tap and the primary resistance Le of 41 mQ result in total copper losses of approximately 0. To compute the temperature distribution in the multilayer. the secondary winding resistance of 0.8 ‘C/W 2. one contact between the multilayer and the ferrite on both sides needs the amount of heat dissipated in each layer as well as the unless each core is grinded to fit each winding package individuthermal resistance of each layer. 117 pF 164 pF 589 pF CPS calc.tory models of a dc-dc converter the ML-PCB transformer ers specifications. HIGH-VOLTAGE INSULATION temperature rise remains moderate. a worst-case thermal analysis layers can be neglected. In the case of heat transfer to one side of the multilayer. In practice.9 OC/W . For this reason. ” and the requirements Due to the low dc resistance of the windings. For this. the cluttering process. DC LS Estrov design (1986) 120 mO 1. category of “distance through insulation.VAN DER LINDE et al. This tolerance applies to the resin layers the thermal conduction in the axial direction.7 mO 450 nH 164 pF 640 pH 74 fs2 Assuming an effective secondary current of 25 A. During this process. Two situations will be discussed: first.25 mil 2200 nH 75 pF 445 WH 165 fs2 Planar transformer 41 mO 0. multilayer. the insulation tion was the reproduction of the parasitic effects. 633 pH meas. Another explanation could be the pres.23 0. however. with the good heat transfer to the environment. primary and secondary windings are sepa.67 “C/W 1. 1. the effect on the represented by an equivalent electrical network of current sources leakage inductance and interwinding capacity lies within a tolerrepresenting the heat dissipated in the copper layers and of ance of ca. Flanagan. vol. Power Electron. pp. 31.113-123. 1127-1132. D. ZEEE Power Electron. 1986. “Planar magnetics for power converters. P. 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